Amplifier having digital micro processor control apparatus

ABSTRACT

This invention relates to an amplifier having digital micro-processor control apparatus and, in particular, to a high frequency power amplifier that includes a micro-processor control system to accurately regulate the operating point of the various amplifying elements in the high frequency power amplifier. The basic amplifier circuitry consists of a micro-controller, a variable voltage attenuator (VVA), a digital to analogue converter and an EEPROM. The EEPROM provides a lookup table which is read by the micro-controller, which then writes to the digital to analogue converter to set the control voltage to the variable voltage attenuator.

FIELD OF THE INVENTION

This invention relates to an amplifier having digital micro-processorcontrol apparatus and, in particular, to a high frequency poweramplifier that includes a micro-processor control system to accuratelyregulate the operating point of the various amplifying elements in thehigh frequency power amplifier.

BACKGROUND TO THE INVENTION

The gain of an amplifier, especially a power amplifier will change atdifferent frequencies, temperatures and operating levels, and betweendifferent units, unless the design compensates for these effects. It isa problem in the field of amplifiers to accurately and dynamicallycontrol the operation of the amplifying elements that comprise theamplifier.

In the case of amplifiers employed in communications systems, fastresponse times are required together with high output powers. Further,the input power and frequency can be different in successive bursts, soa new power level/attenuation must be set, for example in GSMapplications, every 577 μs. This means that the gain control must befast, which limits the amount of processing that can be done in realtime.

A problem typically encountered in amplifiers is that the amplifyingelements, exhibit a fairly significant variation in theircharacteristics due to manufacturing tolerances. In addition, variationsin operating temperature cause a shift in the operating point of theseelements as does ageing of the amplifying elements. It is a typicalprocedure to fine tune the amplifier operation during the amplifiermanufacturing process to compensate for the diversity of manufacturedamplifying elements. Dynamic changes in operating environment can becompensated for by analogue feedback circuitry that is typically foundin an amplifier. This analogue feedback circuitry can provide somerudimentary control over the quiescent point of the amplifying element,although these feedback schemes typically can not compensate forvariation in the operating characteristics of the devices.

In the embodiment shown in FIG. 1, the power amplifier also employs amicroprocessor to change the Quiescent Collector Current (Icq) to threecascaded class AB bipolar power transistors possibly every GSM burst(577 μs) in the case of GSM applications, depending on the poweramplifier output power (GSM base stations use up to 42 dB power controlon each burst depending mainly on the mobile position.). The lcq ischanged to make correction of the transistors' self-heating effectpossible. The power transistor also suffers from an internalself-heating effect; at high powers, the junction gets hotter, and thevoltage applied to the base (Vbe) requirement is less to maintain thesame lcq; this is normal functioning of the class AB transistor and thecollector current (Ic) due to the dc bias goes up with increasing power(This can only be seen by measuring Ic immediately after the rf isremoved.). However, a transistor has a thermal time constant and thereis a transition period between different dissipated powers where the lcqis incorrect for maintaining near constant gain; this has to becompensated for with the bias.

A short time constant heating effect which is observed as rounding ofthe GSM burst edges (known as interburst ripple), and a long timeconstant heating effect which is observed as change in burst gaindependant on the power history (known as history effect). The biasshaping circuit corrects for the interburst ripple with a polarizedshort decay differentiator fed through a very short time constantintegrator. The history effect is corrected by a long time constantintegrator.

One common solution to the first problem is to use a number of trimmerswhich are adjusted during production test to compensate for some ofthese effects individually. Such trimmers may be used to supplement theperformance of a variable voltage attenuator which would be under thecontrol of a micro-controller. The use of trimmers increases the set-uptime, which adds to the cost of the product. If a micro-controller andvariable voltage attenuator are also used then the component coststogether with the costs of setting trimmers and other associatedequipment would be high.

It is possible, in theory to correct for static gain by adjusting theicq to compensate for the gain expansion curves; however, this does notcorrect for the dynamic signal and the history effect and interburstripple effect was observed which made the gain out of specification. Twointegrators with the Icq never going beyond lcqmin and lcqmax can alsobe employed, but this does not fully achieve dynamic gain specification.Additionally, a polarized differentiator was required to fully correctgain for interburst ripple.

U.S. Pat. No. 4,924,191 provides an amplifier arrangement having digitalbias control apparatus that uses a processor to provide precise, dynamiccontrol over the operating point of a plurality of amplifying elementsin an amplifier. This processor controls each amplifying element tooptimise the operating point of each individual amplifying element as afunction of the amplifying element characteristics, the operatingenvironment and the applied input signal. If the measured values of biassignal and output signal do not match predetermined desired values asstored in the processor memory, the processor updates the predefinedbias value that is stored in memory to therefore shift the nominaloperating point of this amplifying element to compensate for dynamicchanges in the operating environment or the operating characteristicsinherent in this particular device. This disclosure sets bias employingdynamic feedback but does not, however, provide a dynamic pulse shapingcircuit.

OBJECT TO THE INVENTION

The present invention seeks to provide an improved apparatus and methodfor controlling a power amplifier. The present invention also seeks toprovide an efficient and inexpensive method of dynamically maintaining aconstant gain in a power amplifier whilst the input power, inputfrequency and temperature are varying. The present invention furtherseeks to provide an amplifier circuit which compensates for variationsin gain due to self-heating effect in a dynamic fashion. The presentinvention further seeks to provide an amplifier circuit whichcompensates for variations in gain with changing output power.

SUMMARY OF THE INVENTION

In accordance with a first aspect of the invention, there is provided anamplifier circuit comprising an amplifier chain, a micro-controller, avariable voltage attenuator (VVA), a digital to analogue converter andan EEPROM, wherein the EEPROM provides a lookup table which is read bythe micro-controller, which is operable to the digital to analogueconverter to set the control voltage to the variable voltage attenuator.The control voltage is updated continually—every 577 μs, for example, ina GSM application. The lookup table can be derived by measuring a fewpoints at room temperature for each power amplifier and then usingpreviously measured typical data to calculate a full lookup table tocompensate each power amplifier. A routine is provided whereby tocompensate the amplifier gain via the variable voltage controller forthe following variables: Unit to unit gain variation; Unit to unitfrequency response; Unit to unit gain variation as a function of backoff; and Unit to unit gain variation as a function of temperature. Theroutine takes data measured from individual amplifiers during productionset up along with known ‘typical’ responses and derives a set of EEPROMcoefficients which are provided to amplifier controllers. The microcontroller uses the EEPROM as a simple look up table to derive variablevoltage controller settings.

The routine comprises of 3 parts: that of data gathering during unittest; that of calculating operating coefficients; and that of runningthe amplifier using the coefficients. The actual calculations involvedin deriving the coefficients are performed in circuitry, for example ina computer outside the amplifier, and can be made as complicated asrequired, since there is no real time constraint on this operation.

In accordance with a still further aspect of the invention, there isprovided a power amplifier which also employs a microprocessor to changethe Quiescent Collector Current (Icq) to some class AB bipolar powertransistors in the amplifier. The lcq is changed to make correction ofthe transistors' self-heating effect possible. This provides a goodefficiency which is obtainable at high powers and nearly constant gainachievable from the transistor at different power levels.

In accordance with a still further aspect of the invention, there isprovided a bias shaping circuit, which uses a polarized differentiatorwith integrator combined with another integrator can then be applied tothe transistor bias. Here, the micro-controller is employed to set Icq,as a linear function of power dissipated. A waveform is thussynthesised, which waveform is shaped with differentiators andintegrators.

In sliding the bias, the output from digital to analogue converter isvaried during each burst. (This voltage controls Quiescent Collectorcurrent, Icq). This allows near constant gain with output power andenables a low Icq at high powers which improves efficiency.

The invention thereby provides an efficient and inexpensive method ofdynamically maintaining a constant gain in an amplifier whilst the inputpower, input frequency and temperature are varying. A relatively fewpoints are measured on each amplifier to provide this compensation fromamplifier to amplifier without the need to adjust trimmers on each unit,thereby reducing production test time. This present invention offers asimple, inexpensive and easy to set up automatic test facilitycorrection method for the production of signal amplification circuitry.The present invention also avails itself to the simplifying ofautomation during the production of amplifiers.

BRIEF DESCRIPTION OF THE DRAWING

In order that the present invention can be more fully understood and toshow how the same may be carried into effect, reference shall now bemade, by way of example only, to the Figures as shown in theaccompanying drawing sheets wherein:

FIG. 1 shows an amplifier line up showing compensation elements;

FIG. 2 details the final cascade of three amplifier stages of FIG. 1;

FIG. 3 shows the stages in an algorithm;

FIG. 4 shows a method of deriving coefficients for the look up table;

FIG. 5 shows a form of an EEPROM look up table;

FIG. 6 shows a GSM communications protocol inter-burst timing diagram;

FIG. 7 shows a typical GSM signal input to a power amplifier;

FIG. 8 shows an output from a power amplifier having the input of FIG.6;

FIG. 9 details one signal transition;

FIG. 10 shows the properties of a class AB bipolar transistor;

FIG. 11 shows a sliding bias circuit of FIG. 12 connected to anamplifier circuit;

FIG. 12 details a bias circuit;

FIGS. 13-16 show output waveform using the circuit of FIG. 11;

FIG. 17 details the derivation of the Vbe waveform;

FIG. 18 shows final and penultimate gain stages;

FIGS. 19 and 20 show the Vbe waveform.

DETAILED DESCRIPTION

There will now be described, by way of example, the best modecontemplated by the inventors for carrying out the invention. In thefollowing description, numerous specific details are set out in order toprovide a complete understanding of the present invention. It will beapparent, however, to those skilled in the art, that the presentinvention may be put into practice with variations of the specific.

In many applications for a power amplifier, the amplifier must maintaina near constant gain over a wide variety of input conditions. A simple,uncompensated amplifier is not capable of meeting the performancerequirements and so some form of compensation mechanism is required.Most TDMA power amplifiers (GSM, DCS1800, PCS1900, IS136) and CDMA poweramplifiers where rapid power control is used with class AB transistorsrequire some form of dynamic correction.

A class A amplifier is one in which the operating point and the inputsignal are such that the current flows at all times in the outputcircuit of the amplifier, whether the collector, plate or drainelectrode of the amplifying element. A class A amplifier operatesessentially over a linear portion of the amplifying elementcharacteristic. A class B amplifier is one in which the operating pointof the amplifying element is at an extreme end of its characteristic, sothe quiescent power is very small and either the quiescent current orthe quiescent voltage is approximately zero. If the input signal issinusoidal, amplification takes place for only one-half a cycle of thesinusoidal input signal. A class AB amplifier is one operating betweenthe two extremes defined for class A and class B amplifiers. Hence theoutput signal is zero for part but less than one half of an inputsinusoidal signal cycle.

Referring now to FIG. 1, there is shown an amplifier circuit. The basichardware consists of a micro-controller, a variable voltage attenuator(VVA), a digital to analogue converter and an EEPROM. The EEPROMprovides a lookup table which is read by the micro-controller, whichthen writes to the digital to analogue converter to set the controlvoltage to the variable voltage attenuator. The control voltage isupdated continually—every 577 μs, for example, in a GSM application. Thelookup table is derived by measuring a few points at room temperaturefor each power amplifier and then using previously measured typical datato calculate a full lookup table to compensate each power amplifier.

For production testing the variable voltage attenuator can be adjustedto provide the correct gain at a few input levels and frequencies atroom temperature and the corresponding digital to analogue convertervalues are stored. An interpolation routine is used to calculate theother room temperature values to reduce the number of points measured,keeping the test time short.

Following this, a few measurements are determined to characterise thenon-linear response of the variable voltage attenuator. A typical curveis fitted to these points to reduce the number of points measured,keeping the test time short. The typical gain response of theuncompensated power amplifier versus temperature has beenmeasured/derived in the lab and this response is combined with the abovemeasurements.

Since a variable voltage attenuator response is non-linear the gainversus temperature is measured in dBs, as opposed to digital to analogueconverter steps because the number of digital to analogue convertersteps varies from power amplifier to power amplifier. The variablevoltage attenuator response is used to convert the room temperaturedigital to analogue converter values into dBs, which are combined withthe gain vs. temperature points. Then the inverse variable voltageattenuator function is used to convert these values back into digital toanalogue converter values which are stored in the EEPROM.

The micro-controller also implements a simple interpolation routine toimprove the resolution of the compensation with respect to temperature,without increasing the size of the lookup table. The routine was keptvery simple by carefully matching the step sizes of the variable voltageattenuator and the temperature sensor.

In one aspect of the invention there is provided a routine which canalso act as a gain compensation algorithm, which can be used to derivethe EEPROM coefficients and how these should be interpreted by themicro-controller.

The routine is intended to compensate the amplifier gain via thevariable voltage controller for the following variables:

1) Unit to unit gain variation

2) Unit to unit frequency response

3) Unit to unit gain variation as a function of back off

4) Unit to unit gain variation as a function of temperature.

The routine takes data measured from individual amplifiers duringproduction set up along with known ‘typical’ responses and derives a setof EEPROM coefficients which are provided to amplifier controllers. Themicro controller uses the EEPROM as a simple look up table to derivevariable voltage controller settings.

The routine comprises of 3 parts: that of data gathering during unittest; that of calculating operating coefficients; and that of runningthe amplifier using the coefficients. The actual calculations involvedin deriving the coefficients are performed in circuitry, for example ina computer outside the amplifier, and can be made as complicated asrequired, since there is no real time constraint on this operation.

An object of the routine is to separate temperature out from back off(output power) and frequency response as an orthogonal variable. This isbecause output power and frequency response will be characterised foreach unit in production, whilst temperature response will not.Therefore, there is a requirement to be able to add temperaturecompensation based on a ‘typical’ response.

The coefficients are calculated by deriving a set of mathematicalfunctions fitted to the measured data, as defined by the following stepswith reference to FIG. 3.

Step 1: DAC=f1(Po,freq)

From production measurements, the values of DAC required to maintain aconstant desired gain as a function of frequency and output power aredetermined. For this test, the required input power is set (=requiredoutput power−required gain) then DAC is incremented until the requiredoutput (Po) is achieved.

Typically, in a production environment, it would be possible to restrictthe number of measurements required by assuming that the gain isconstant as a function of Po below a power level corresponding to classA operation (TBC) and by restricting the number of frequency points.Using a mathematical surface fit routine, the closed form function forDAC as a function of Po and freq is derived. It may be necessary tointerpolate between freq and Po points in order to obtain a matrixsuitable for surface fitting.

An assumption made here is that a nearly constant amplifier temperatureis maintained during the testing. Self heating effects are discussedlater on.

Step 2: A=f3(DAC,freq)

In step 2, a corresponding gain numbers (in dB) to the DAC value isobtained. This step is required as variable voltage controller responsesare not perfectly linear or repeatable from unit to unit. This steptherefore calibrates out variable voltage controller responsevariations.

From production measurements at low power, the amplifier gain ismeasured as a function of DAC and freq. It is important here that theamplifier remains in class A even when the variable voltage controlleris at maximum gain. The optimum number of readings should be small asthe required function is typically smooth and well behaved. Using asurface fit routine, the closed form function for A as a function of DACand freq is derived.

Step 3: A′=A+ΔA

This step adds a gain compensation factor which is derived later. It isa simple addition function:

Step 4: DAC′=f4(A′,freq)

Using surface fitting on the same measurement data points as used forstep 2, the inverse function to f3 is derived as a closed form function.It is possible to identify errors caused in the curve fitting processand digital to analogue converter resolution at this stage by runningthe two functions f3, f4 back to back with ΔA set to zero.

The output from step 4 is now the required compensated variable voltagecontroller digital to analogue converter value.

Step 5: Tn=kn*ADC(Tn)+cn

Step 5 calibrates the temperature sensors. All of the temperaturesensors rely on the linear variation of the Vbe of a transistor withtemperature. Depending on the sensor, this Vbe value can be amplifiedand/or inverted.

The coefficients (k1, k2, k3) are pre-characterised and is convenientlyassumed as constant for all units. Offset values (c1, c2, c3) arederived in the production test of each unit at room temperature. If thetemperature response of the amplifier is fairly linear, as expected,then there is no need to relate T1, T2, T3 to absolute temperatures.These values may, therefore, be calculated with respect to factoryambient (T=0=steady state ambient whereby, cn becomes −kn*ADC(Tn) atambient).

Step 6: ΔA=f2(Tn,Po)

In step 6, the temperature compensation in dB, derived byexperimentation is calculated.

The micro-controller is employed to set a variable voltage attenuator(VVA) to maintain the overall gain. The micro-controller sets thevariable voltage attenuator based upon a lookup table which iscalculated and loaded during Production Test.

The Input Power is permitted a particular range, as follows:

Static Attenuation = 0 . . . 12 dB (in 2 dB steps) 3 bits DynamicAttenuation = 0 . . . 30 dB (in 2 dB steps) 4 bits

The range is accordingly equal to 0.42 dB (which is equivalent to 22values) If the static attenuation is incorrect, e.g. equal to 14 dB,then DAC_MAX_IP levels will be used. The frequency figure is a 7 bitnumber, 128 values in 1.6 MHz steps. The use of 16 steps provides 4.8MHz resolution in the 75 MHz operating band.

The temperature can be measured via the analogue to digital converterfrom three sources. The routine only uses the sensor on the ControlBoard. Table 1 shows the position and approximate resolution of eachsensor. All 3 temperature sensors are read using 10-bit analogue todigital converter resolution, and are then converted into an 8-bitnumber range by subtracting the offset shown in the last column. Forexample temperature sensors 2 and 3 both have an input voltage range ofapproximately 1.25V to 1.6V, which corresponds to an approximateanalogue to digital converter reading of 260 to 330. Subtracting 200means the reported reading ranges from 60 to 130 which can berepresented by an 8-bit number.

The variable voltage controller lookup table is conveniently stored inthe parallel EEPROM, which is a 3-dimensional array of digital toanalogue converter value versus input power, frequency and temperature.Typically, the EEPROM will store values for all 22 input powers, for 16frequency values and for 17 temperature values, providing a table of5984 bytes (22×16×17).

The values for the control levels are determined every burst. Thecalculation uses information from a modulator and therefore thecalculation cannot start until at least the 2^(nd) byte has beenreceived from the DRX. In this case three bytes are sent every burstproviding data relating to the carrier attributes as well as check data.

To increase the accuracy of the compensation without increasing the sizeof the lookup table, ‘simple interpolation’ is carried out between thetemperature values. The temperature values are stored at 5° C. intervalsand the gain decreases by approximately 0.5 dB for every 5° C. increasein temperature. The average variable voltage controller step size isapproximately 0.13 dB. Table 2 shows an example of how this works. Ifthe analogue to digital converter value is 150 or 145 the value is takendirectly from the lookup table in the EEPROM. Between these two values 1or 2 digital to analogue converter steps are added to or subtracted fromthe closest value in the lookup table.

If the temperature is below 0° C. or above 80° C. then the closest valuefrom the lookup table is used, but no further ‘interpolation’ is carriedout. The analogue to digital converter value corresponding to roomtemperature is programmed into the lookup table and the code assumesthat the temperature will be 10° C. higher for every step the analogueto digital converter is below this value. Room temperature is theanalogue to digital converter value that is read during the productiontest characterisation routine. Between bursts the control levels will beset to the new levels, which may be the same as the previous one. Thedigital to analogue converters are written to every burst even if thecontrol level has not changed.

The timing of the data transfer to the 4 digital to analogue convertersis shown in FIG. 6. The variable voltage controller is changed as soonas possible after the end of the burst plus 10 μs. It is delayed by 10μs because the burst could still be at full power up until this time.Therefore the variable voltage controller is changed after about 12 μsand the three bias levels are set just before the variable voltagecontroller is changed.

In the embodiment shown in FIG. 1, the power amplifier also employs amicroprocessor to change the Quiescent Collector Current (Icq) to threeclass AB bipolar power transistors possibly every GSM burst (577 μs),depending on the power amplifier output power. (GSM basestations use upto 42 dB power control on each burst depending mainly on the mobileposition.) The lcq is changed to make correction of the transistors'self-heating effect possible, good efficiency obtainable at high powersand nearly constant gain Achievable from the transistor at differentpower levels. The actual value of lcq follows a curve based on the powerdissipated within the transistor and not just compensating for gainexpansion. This ensures that the correction for transistor self-heatingworks during dynamic burst (see below), in addition to achievingadequate static flat gain with power backoff. FIG. 2 shows in greaterdetail the micro-controller's interaction with the last three cascadedamplifiers.

In order to overcome the internal self-heating effect at high powers,use is made of a thermal time constant associated with the transistorand a transition period between different dissipated powers where theIcq is incorrect for maintaining near constant gain, and which iscompensated for with the bias. This is carried out by integrating anddifferentiating the bias voltage simultaneously and applying thecomposite signal to the transistor bias. A further feature of themicro-controller is that it can be employed to change the Icq withpower. A bias shaping circuit, which uses a polarized differentiatorwith integrator combined with another integrator can then be applied tothe transistor bias.

The stages of operation are as follows: The micro-controller is employedto set Icq, as a linear function of power dissipated. A waveform is thussynthesised, which waveform is shaped with differentiators andintegrators.

Referring now to FIG. 6, there is shown inter-burst ripple (edgerounding) and history effect (a low power burst after a high power bursthas increased output gain). Both of these effects are caused by thethermal lag in the transistor. When the power is changed abruptly (5μs),the transistor die takes considerably longer to cool down or heat up tothe temperature which would be observed in a static power level case.

In sliding the bias, the output from the digital to analogue converteris varied during each burst. (This voltage controls Quiescent Collectorcurrent, lcq). This allows near constant gain with output power (cf FIG.9) and enables a low lcq at high powers which improves efficiency.Further, this drives the sliding bias circuit which corrects forproblems in FIG. 6. Sliding bias without the pulse shaping circuit ofFIG. 12 in this way solves the issue of constant gain and permits theuse of a low lcq at high powers to improve efficiency. FIG. 17 shows theVbe waveform. Adding the pulse shaping circuit has the desired effect ofvarying Vbe to compensate for the thermal time-constant in thetransistor.

Referring to FIG. 18, the sliding bias digital to analogue converteroutput can be varied over temperature as well as power to compensate forgain change with temperature in final stages. As temperature increases,P3 remains constant but P2 increases, due to gain change in Q3.Therefore power dissipated in Q2 varies with temperature and poweroutput. A look-up table array can be used to store required bias value,as shown in table 3

Note that the differentiator is polarised to protect the transistor:Transistors are very sensitive to Vbe; just 10 mV too much can destroy adevice. Referring to FIGS. 19 and 20, the spike shown by the dotted linein FIG. 20 could destroy the transistor in some cases.

The problems solved by the present invention are, inter alia, thecorrection of gain due to the self-heating effect (dynamic correction),the maintenance of gain at an effectively constant level with changingoutput power (static correction), and the maintenance of good efficiencyat high powers. Most TDMA power amplifiers (GSM, DCS1800, PCS1900,IS136) and CDMA power amplifiers where rapid power control is used withclass AB transistors require some form of dynamic correction. Thisinvention offers a clever, simple, inexpensive and easy to set up on anautomatic test facility correction method.

What is claimed is:
 1. An amplifier circuit having an amplifyingelement, a bias circuit coupled to an input of said amplifying element,and a microcontroller arranged to control the bias circuit according topredetermined criteria and determined input power to the element inorder to control the amplifying element gain in order to compensate forthe internal heating effects of said amplifying element, thecompensation taking into account thermal lag in the amplifying elementto provide more accurate control during dynamic changes in thermaleffect.
 2. An amplifier circuit according to claim 1 wherein themicrocontroller comprises a memory device and the predetermined criteriaare a number of bias control settings stored in the memory whichcorrespond to predetermined input power values such that themicrocontroller controls the bias circuit according to the settingcorresponding to the determined input power.
 3. An amplifier circuitaccording to claim 1, wherein the amplifying element gain is furtherdependent on: measured temperature; input frequency; specific batchdependent characteristics of the amplifying element; a combination ofthese.
 4. An amplifier circuit according to claim 1, wherein theamplifying element is a bipolar transistor.
 5. An amplifier circuitaccording to claim 2, wherein the memory device is an EEPROM.
 6. Anamplifier circuit according to claim 4, wherein the microcontroller isarranged to control the bias circuit in order to control the QuiescentCollector Current (Icq) dependent on output power of the amplifiercircuit.
 7. An amplifier circuit according to claim 2, wherein themicrocontroller is arranged to implement an interpolation algorithm formeasured values of input power between those stored.
 8. An amplifiercircuit according to claim 2, wherein the bias control settings arederived from previously measured operational parameters of the amplifiercircuit.
 9. An amplifier circuit according to claim 1 wherein thedetermined input power is measured in regular time slots.
 10. Anamplifier comprising the amplifier circuit according to claim 1 andfurther comprising a gain control device coupled to an input of theamplifying element in order to control the input power to the amplifyingelement, the gain control device being controlled by the microcontrollerdependent on measured input power to the amplifying element.
 11. Anamplifier according to claim 10 wherein the gain control device is avariable voltage attenuator.
 12. An amplifier comprising the amplifiercircuit according to claim 3 and further comprising a variable voltageattenuator coupled to an input of the amplifying element in order tocontrol the input power to the amplifying element, the attenuator beingcontrolled by the microcontroller dependent on the determined inputpower to the amplifying element.
 13. An amplifier comprising a number ofamplifying circuits according to claim 1 each of said amplifyingcircuits comprising different predetermined criteria.
 14. A method ofoperating an amplifying element in order to compensate for internalheating effects for different input signal levels, said methodcomprising: determining input signal level to said amplifying element;and controlling bias conditions of said element dependent on said inputlevel and predetermined criteria while taking into account thermal lagin the amplifying element to provide more accurate control duringdynamic changes in thermal effect.
 15. A method according to claim 14wherein said predetermined criteria are a number of bias controlsettings which correspond to predetermined input power values such thata microcontroller controls the bias conditions according to settingcorresponding to determined input power, and/or an algorithm which isdependent on the determined input power.
 16. A method according to claim14 further comprising controlling the input signal level to theamplifying element dependent on the determined signal input level.